Large signal VCO

ABSTRACT

An alternation voltage- or current generator comprises a first switch driving output network whose frequency can be tuned. The tuneable network comprises a first Inductor that is coupled with a first capacitor. A second inductor and/or at least a second capacitor and/or at least a series circuit of a third inductor and a third capacitor which is coupled via at a second switch to the network. The second switch is controlled by a controlled delay (PWM) which is synchronized by a sign change of current and/or voltage in the network.

FIELD OF THE INVENTION

The present invention relates to the generation of electromagnetic wavesby means of resonant network and more particularly to near field powertransmission for RFID and wireless power supplies. Furthermore, thecurrent invention relates to control the resonant frequency of anoscillator by means of a switched resonant network.

BACKGROUND OF THE INVENTION

Wireless power supplies can be realized by means of inductive and/orcapacitive near-field coupling. This is used in many RFID systems andwireless battery chargers. A source unit, hereinafter referred to as thebase station, generates an electromagnetic alternating field whoseradiation unit represents a resonant circuit. By means of such aresonant circuit the alternating field is produced efficiently atrelatively low cost, because the driver circuits generate virtually noswitching losses (ZVS zero voltage or zero current switching ZCS). Thefiltering effect of the resonant circuit influences the spectral powerdensity and suppresses harmonics. A generator in the base station drivesthe resonant circuit with a signal whose frequency and amplitude can bevaried to control the transmitted power.

The designated energy equipment to be supplied, hereinafter referred toas the load unit is placed at a distance of a fraction of the wavelengthof the alternating electromagnetic field so that the coupling conditionbetween the base station and the load unit becomes optimal. This isachieved by the same polarization, minimum distance and greatestcoupling surface. Often, additional soft magnetic materials are used inorder to further increase the coupling and/or to direct the fieldprofile.

It is known, that the transmitted power at a given coupling factor canbe increased by the use of a resonant circuit in the load unit.Optimally, the load resonant circuit shall be tuned exactly to thefrequency of the generated alternating magnetic field, where one has toweight achievable coupling gain versus resulting bandwidth.

In “System description Wireless Power Transfer”, a resonant circuit isshown, which is driven by a generator.

Therein, in a resonant circuit a plurality of inductances are added byswitches to the resonant circuit to concentrate the radiated fieldenergy to an area where load units are positioned. Additionally, anupstream voltage or current regulator controls the power supply.

The general disadvantage of wireless power transfer principles usingresonant circuits in the base station and/or in the load unit is theresonance frequency detuning. This has been due to component tolerances,component aging, variable coupling and load conditions. This loaddetuning is undesirable because the impedance of the resonant circuit isfrequency selective and a predetermined operating frequency no longermatches the resonant frequency. Consequently the overall efficiency ofthe driver circuit decreases and thus the power transmission from basestation to the load unit. In addition the drive waveforms become moredistorted and the driver circuits generate more harmonics. Thewell-known network measurement method measures while in an interval, theresonant frequency of the network and then it operates at thisfrequency, but has no ability to control the network frequency actively.This would be very desirable, because regulations such as EN300330,ITU-REC7003 and RSM2123 specify maximal amplitude values for givenfrequency ranges (for example 119 . . . 135 kHz). Furthermorecountry-specific narrow band frequencies exist within a given frequencyrange which define much smaller limiting values.

Therefore, it is important to control not only the power but also thespectral position of the radiated power.

U.S. Pat. No. 6,586,895 shows how to control an inductor or a higherorder network using a variable coupling interval during both half wavesof a resonant circuit period. FIG. 1 shows the main circuit wherein thecapacitor CS and the inductance LH form a series resonant circuit. Thecapacitor CM and the inductor LM are both coupled via the controlledtransistors Q1 a and Q1 b and their integrated body diodes to CS in twopart-intervals of the resonant circuit period. The coupling controloperates in both half cycles of the resonant circuit period, since thecurrent in CM and LM depends in both directions from the control of thetransistors Q1 a respectively Q1 b. The capacitor CR is not relevant,because it remains short-circuited by the transistor QH and diode DH, orthe transistors Q1 a and Q1 b remain fully open or fully closed when QHis open.

In “Stabilisation the Operating Frequency of a Resonant Converter forWireless Power transfer to Implantable Biomedical Sensors” a controlledresonant network is proposed in a generator, wherein the bad resonantcircuit can be controlled by a variable coupling interval (see FIG. 2a). In an LC resonant circuit, a second capacitor is coupled forsub-intervals of the resonant circuit period to the LC resonant circuit(see FIG. 2 b). The switch control compares the resonant circuit voltageversus a reference voltage which acts as a control variable and thecomparator output represents the switching signal for the couplingswitches (see FIG. 2 c). This control method works very unreliable,since any change in the resonant circuit amplitude immediately interactwith the frequency of the resonant circuit.

Coupled load units (respectively changing load conditions) alter theamplitude of the resonant circuit voltage. Also, often used amplitudecontrol methods to control the power transfer, always result infrequency detuning. In fact, the amplitude control and the frequencycontrol interact with each other, they cannot be controlledindependently.

Generating control signals, which are dependent on the controlled outputsignal is very difficult. The main problem is, that an updated outputvalue instantly changes the input value and thus makes the systemunstable.

Another important requirement of a base station, whose frequency can becontrolled, is the amplitude fidelity. The amplitude shall remainconstant, when the frequency changes. This is especially difficultduring the sudden step tuning (frequency hopping), because the impedanceof the resonant circuit may vary greatly according to the frequencytuning step size.

In a tunable resonant circuit using controlled interval coupling for itscomponent variation, it is very vital that the coupling interval can becontrolled independently from the resonant circuit period andamplitudes.

This means the resonant circuit frequency is only a function of acontrol variable. This independence condition ensures stable operationregardless of the quality of the response network. Frequency or phaselocked loops should preferably have a short lock time. Thereby,frequency or phase changes in the network are immediately corrected. Inaddition, a fast control loop is important to manage in a sweepoperation an operating frequency range with large time resolution (shortresidence time).

The following invention describes a method and their detailedimplementations to control the frequency or phase of a generator viacontrolled large signal network. The following invention describes amethod that meets all requirements above and is characterized by a shortlock time, quality independent stable operation and high efficiency.

SUMMARY OF THE INVENTION

In a first aspect of the present invention, a wireless power supplycircuit is proposed, which controls the resonance frequency of an LCresonant circuit electronically. A capacitance, inductance or combinedarrangements of both are coupled by means of electronic switches to thenetwork during a part interval of the resonant circuit period. A pair ofadditional switches is coupled to the LC resonant circuit to generate acontinuous wave.

In another aspect of the invention, the pair of switches, whichgenerates a continuous wave in the resonant circuit, forms anoscillator.

SHORT DESCRIPTION OF DRAWINGS

FIG. 1 shows a circuit to control the frequency of a large signalnetwork according to a first prior art.

FIG. 2 a . . . 2 c show circuits to control the frequency of a largesignal network according to a second prior art.

FIG. 3 shows a block diagram of a large signal VCO according to a firstexemplary embodiment of the invention.

FIG. 4 shows a block diagram of a controlled large signal capacitoraccording to the FIG. 3.

FIG. 5 shows waveforms according to FIG. 4.

FIG. 6 shows a block diagram of a large signal VCO according to a secondexemplary embodiment of the invention.

FIG. 7 shows a block diagram of a controlled large signal inductoraccording to the FIG. 6.

FIG. 8 shows waveforms according to FIG. 7.

FIG. 9 shows a block diagram of a large signal VCO according to a thirdexemplary embodiment of the invention.

FIG. 10 shows a detailed block diagram according to FIG. 9.

FIG. 11 shows waveforms according to FIG. 10.

FIG. 12 shows a detailed circuit diagram of an exemplary embodiment ofthe invention.

FIG. 13 shows a detailed circuit diagram of an exemplary embodiment ofthe invention.

FIG. 14 shows a detailed circuit diagram of an exemplary embodiment ofthe invention.

FIG. 15 shows a frequency management method according to the invention.

FIG. 16 shows an expanded frequency management method according to theinvention.

DETAILED DESCRIPTION OF THE INVENTION

The block diagram of a series push-pull oscillator with a controlledcapacitor in the resonant network shown in FIG. 3 according to a firstembodiment. Herein, this oscillator is used as a base station in awireless power supply system, but it can be used as any oscillator. Theswitches Qv1 and Qv2 form a series push-pull stage (half bridge), whichcouples its center alternately either to VCC′ or to a referencepotential (ground). The switches Qv1 and Qv2 are alternately open orclosed. In one embodiment, the switches Qv1 and Qv2 are of the samechannel type (either two P-channel or two N-channel MOSFETs respectivelyeither two PNP or two NPN IGBTs). The opposite-phase drive signals V1and V2 are guaranteed by the inverter (Inv). In another embodiment, theswitches Qv1 and Qv2 are of complementary types (P/N channel MOSFETs orPNP/NPN bipolar transistors or IGBTs), there Inv is not implemented. Thecenter tap of the switches is connected to a series resonant circuitconsisting of one or more inductors L01 . . . L0 n and one or morecontrolled capacities Ctot with a reference potential (ground). At leastone inductor L01 is part of the resonant circuit. If additional multipleinductors are used, any number of them are coupled via switches Q01 . .. Q0 n to the resonant circuit. In any arbitrary branch Q0 n, L0 n maybe additional series inductors and their switches (not shown). Theseswitches control the series inductor bypass. Such extended inductorcircuitries may be required at high VCC′, higher transmitted powerand/or multiple load units.

A load unit comprises a load resistor (Rload), a pickup inductor (Lsec)and often, but not necessarily a capacitor Csec, that tunes the loadunit's resonant frequency to match with the base station radiationfield. Alternatively, a parallel resonant circuit can also be used inthe load unit (not shown). The proper tuning to either series- orparallel resonance in the load unit is not critical, because theattenuation of Rload the increases the bandwidth significantly.

All or any number of the inductors L01 . . . L0 n are implemented as anopen coils and couple one or more load units to the resonant circuit ofthe base station. If multiple inductors L01 . . . L0 n are used, theyare positioned on a surface or on multiple surfaces which are at anangle to each other. Is the corresponding coupling switch Q01 . . . Q0 nclosed, the current in the corresponding inductance L01 . . . L0 ngenerates an induction vector. Thus, the induction field can bedistributed arbitrarily on a given area or a confined space.Advantageously in this way, the radiated induction field is concentratedon the geometrical positions of the load units. Therefore, merely thoseinductors L01 . . . L0 n are coupled to the resonant network, whichtransferred power to the load units. If only one load unit is present,its coupling with the base station can be improved by coupling moreinductors L01 . . . L0 n to the resonant network. Alternatively, in oneembodiment, only one permanent wired inductor L01 is used in theresonant network of the base station for power transmission. In the nearfield, the arrangement can be regarded as an open transformer whosecoupling (coupling k) strongly varies with the geometric positioning andthe distance from L01 . . . L0 n to Lsec. A feedback closes a feedbackloop from the resonant network to the switch input V1 respectively V2.

At a pole of the impedance, defined by L01 . . . L0 n, Ctot and thecoupled proportion of Csec and Lsec, the resonant circuit develops azero degree phase shift between voltage and current. The switch and theFeedback block develop in their over all transfer function also zerodegree. Consequently, the phase condition for a sustained oscillation isfulfilled. The amplitude condition is also fulfilled, because theresulting attenuation Rload is less than the gain in the loop (Qv1, Qv2,resonant circuit and feedback).

The block feedback taps off a part current from the resonant network andgenerates a square wave drive signal for V1 respectively V2. This drivesignal is further coupled as the signal fist to a phase locked loop(Phase Locked Loop PLL). Therein a phase comparator (Phase Comp.)compares fist versus a reference frequency (fsoll) and filters theresulting error signal in the loop filter. The filtered error signalcontrols the capacitance of the resonant circuit Ctot. The concept ofvoltage-controlled oscillator (VCO Voltage Controlled Oscillator) isused here and in the following as a general term. Here, no distinctionis made whether it is a current- or voltage control because both can betransformed into each other. If the oscillator does not oscillate withfsoll, or detunes the reactance in the oscillator, an error signal isdeveloped at the output of the phase comperator (phase Comp.). Thefiltered error signal adjusts the capacity Ctot until the oscillatorfrequency (fist) becomes equal to the reference frequency (fref). Inthis manner, different inductor values in L01 . . . L0 n, changes in theload unit, component value changes and varying coupling conditions arecompensated. The integrating portion of the PLL controls the residualerror always to zero. The PLL lock time is determined by the responsetime of Ctot and the stability in the PLL. In the current invention Ctothas a response time of half a resonant circuit period. Thus, the PLL isdimensioned that its lock time becomes minimal possible after an unlockcondition. This favors PLL-based synchronization principles versussoftware implementations for such applications because of the smallhardware costs and its shortest lock time. This is because the phasecomparator inside the PLL generates for each network period at least oneerror signal which influences the control loop. A PLL lock time (stepresponse to a fsoll change) of less than 100 oscillator frequency cycleswith a 2nd order low pass filter (loop filter) are easily achieved,unless a frequency prescaler is used in the oscillator signal feedback.Software-based PLL control loops with the same properties can only beachieved with considerably greater effort, because all PLL calculationsshall be done in real time within one oscillator period (calculation ofphase error and low pass filtering).

The operating voltage VCC′ is controlled by pcont in block VCC control.In this manner the oscillator—respecively the resonant networkcurrent/voltage amplitude is controlled to the desired energy level ofthe emitted induction field. Coilselect controls the switches Q01 . . .Q0 n and corresponds to a selection signal of the inductors L01 . . . L0n, which are coupled to the resonant network. Coilselect controls thegeometric induction field distribution.

The block Radicontrol generates the necessary signals pcont, coilselect,fsoll and loopselect. The control signal radiationdata includesparameters which are required for generating pcont, coilselect, fsolland loopselect.

Radiationdata containing a first data group, which is required for thegeneration of pcont. This first data group defining pcont directlyand/or are data defining how pcont varies over time. Radicontrolgenerates based on a first value of the first data group a controlsignal (Pcont) for VCC control which generates pulses on VCC′ andaccordingly enables a burst operation in the oscillator. Further,Radicontrol generates based on another value of the first data group acontrol signal (Pcont) for VCC control which control the amplitude ofVCC′ and accordingly, the radiated field energy can be adjustedcontinuously.

Radiationdata contains a second data group which is required for thegeneration of coilselect. This second data group defining coilselectdirectly and/or are data defining how coilselect varies over time.

Radiationdata contains a third data group which is required for thegeneration of fsoll. This third data group defining fsoll directlyand/or are data defining how fsoll varies over time.

Radicontrol generates the signal loopselect based on the signalcoilselect and also based on fsoll if an fsoll update exceeds athreshold (for example 10%). Loopselect modifies the PLL loop gain inthe loopfilter, e.g., by modifying the charge pump current in the outputof the phase comparator. If, e.g., only one inductor L01 is selected bycoilselect, loopselect generates a lower charge pump current,consequently the Loopfliter gain is lower. If several inductors L01 . .. L0 n are selected, the process is reversed and loopselect increasesthe charge pump current, then the Loopfilter gain is higher. Thisresults in a constant PLL loop gain and guarantees uniform dynamic PLLproperties. In a further embodiment of the present invention loopselectconnects an additional capacitance to Ctot. This extends the tuningrange of the VCO and/or switches to different frequency bands.

By means of these three data groups, Radicontrol can generateindependently the three independent transmission parameters: radiatedfield energy, the area or the space of the radiated field and thefrequency of the radiated field.

Thus, e.g., the radiated field energy can be controlled by Pcont at anyspectral position (fsoll) and at an arbitrary spatial position. A simplemodification of radiationdata maps another radiation properties. Thus,e.g., in this way different country-specific limiting field emissionspectra can be generated easily. For this purpose, the third data groupof radiationdata includes data causing fsoll to generate no signal atpredefined spectral positions.

In one embodiment, Radicontrol alters the reference frequency (fsoll)continuously (sweep) and in another embodiment stochastically, in orderto distribute the induction field in a wider spectral band, implementedby excluding predefined fsoll frequencies, spectral positions where nofield levels shall be emitted. Advantageously, this reduces the peaklevel in the frequency spectrum and the over all transmitted power ishigher until a legal standardized maximum is reached. As anotheradvantage, other users using the same frequency range are less disturbedand a given frequency band can be better shared. Further, multiple userscan also negotiate their frequency range minimize interference andjammer. The major advantage of frequency control loops usingphase-locked loops (PLLs) is the independence of the oscillator signalamplitude. This oscillator (VCO) operates always at its ideal conditions(phase and amplitude condition). Consequently, the radiated fieldremains constant at frequency hop transients. FIG. 4 shows a blockdiagram of a controlled large signal capacitor (Ctot(control)) accordingto FIG. 3. Signal waveforms of FIG. 4 are shown in FIG. 5 and are usedin the further description with its indices. The oscillator is connectedat the ports 1 and 2. The inductor L01 (shown as one of a pluralityinductors L01 . . . L0 n) and the capacitor C0 form a series resonantcircuit. The capacitor CM is coupled to C0 via coupling switches Q1 aand Q1 b. The plotted diodes indicate that the coupling switches arecontrolled in only one direction. In the opposite current direction Q1 aand Q1 b remain bridged by the diodes. In the further description, theterm switch defines the functional active controllable part of Q1 a andQ1 b. The term coupling switch defines the functional activecontrollable part and the diode. The first limiting conditions of theswitch control is the one when the coupling switches Q1 a and Q1 b arealways open. Then, the resulting total capacity is equal to C0 andtherefore minimal. Consequently, the resonance frequency is maximal. Thesecond limiting conditions of the switch control is the one when thecoupling switches Q1 a and Q1 b are always closed. Then the resultingtotal capacity is equal to the parallel circuit of C0 and CM andtherefore maximal. Consequently, the resonance frequency is minimal. Forpart interval coupling of CM with C0 any desired intermediate valuebetween the two extreme values can be set for total capacity by thecontrol input Control. Part interval coupling of C0 and CM means acoupling interval defined by a portion of the resulting resonant circuitperiod. This represents a time- or angle interval in the followingdescription. The whole circuit represents a total capacity (Ctot), whosevalue is controlled by Control. The resonant circuit frequency equalsthe average of the entire resonant circuit period. The actual resonantcircuit frequency changes within the resonant circuit period betweenminimum and maximum respectively maximum and minimum based on the statesof the two coupling switches. The resulting capacitor voltage V0 issensed in block V-Sense and coupled to block Sign Detect for signdetermination. The capacitor voltage sensing can comprise a voltagetapping, thereby it is only important that the sign of the output signalhas a defined phase relationship to the voltage V0 across the capacitorC0. The resulting waveform is shown in A. The following differentiator(Diff) derives A after the time and outputs its absolute value, as shownin trace B. This results in positive pulses at each zero crossinginstants 0, T3 and T6 of the resonant circuit voltage V0. The pulsewidth modulator (PWM) generates a sawtooth C (not shown in FIG. 4, sinceit is a signal inside the PWM modulator), which runs synchronously withthe pulses B. This sawtooth trace C is compared versus the control valueControl inside the PWM block. Is Control lower than C, the PWM output Dis high. Is Control greater than C, the PWM output D is low. Based onthe known pulse width modulation (PWM) principle the PWM-modulator canbe realized in various ways. It is only essential that the PWM signal Dis triggered by B and the pulse width is controlled by the inputControl. The subsequent Demux block uses signal A to selects a portionaccording to the corresponding half-wave and couples it to the switchesQ1 a and Q1 b. Is A high, the negated signal D controls the switch Q1 avia E. Is A low, the negated signal D controls the switch Q1 b via F. IsD not linked by A with the output E respectively F, the correspondingoutput E respectively F remains high. The switch Q1 a respectively Q1 bis closed (ON) if E respectively F is high. If E respectively F is low,Q1 a respectively Q1 b is open.

This keeps the switches always at least a half cycle closed and let thembypass their internal diodes. This results in lower conduction lossesand prevents the formation of turn-on transients in the diodes, wherebymay still flow a current in the diodes. This process reduces in any caselosses, especially if the resistance of the switches (Rdson) issufficiently small. When E becomes low, the coupling switch Q1 a opensand it develops a positive half sinusoidal voltage wave in the sum ofVQ1=VQ1 a+VQ1 b during the time interval from T1 to T2. During thisinterval, no current flows through CM. When the voltage VQ1 becomes zero(time instant T2), the stored energy in L0 generates a negative currentflow through the internal diode of the coupling switch Q1 a. This lastsfrom T2 to T3 and is replaced by the above described process ofconducting (or inverse conducting) switch Q1 a. The coupling switches Q1a and Q1 b remain closed until the signal F goes low. It develops anegative half sinusoidal voltage wave in the sum of VQ1 during the timeinterval from T4 to T5. During this interval, no current flows throughCM. When the voltage VQ1 becomes zero (time instant T5), the storedenergy in L0 generates a negative current flow through the internaldiode of the coupling switch Q1 b. This lasts from T5 to T6, and isreplaced by the above described process of conducting (or inverseconducting) switch Q1 b. The entire current flow interval of theinternal diodes is reduced to the interval from T2 to T3 and T5 to T6.In FIG. 4 the capacitors C0 and CM might be connected in series iflarger inductor values and/or a higher operational voltage VCC′ aredemanded (not shown). In this case, the block V-Sense senses the totalvoltage across both capacitors (C0 and CM). All other blocks and theiroperation may remain unchanged. This is also the case if in the seriescircuit of C0 and CM additional capacitors are coupled in parallel tothe switches Q1 a and Q1 b. It is further apparent that the function ofthe block diagram in FIG. 4 does not change when CM is divided into twopartial capacitors and each of the switches Q1 a and Q1 b couples one ofthese part capacitors to C0. Advantageously, then, the two switches Q1 aand Q1 b are in parallel to each other, resulting in smaller conductionlosses.

An oscillator also remains functional when instead of a controlledcapacitor, a controlled inductor is used to control the frequency.According to a further embodiment a parallel push-pull oscillator isshown in FIG. 6, which delivers compared to the half-bridge with thesame circuitry consumes more output power. A symmetrically tapped coil(Ltap1 and Ltap2) is connected at both ends via a switch (Qv1 and Qv2)alternately to a reference potential (ground). Across the two ends ofLtap1 and Ltap2 one or more controlled inductors (L0) and one or moreadditional inductors L01 . . . L0 n are coupled to Ltap1 and Ltap2. Thisis alternatively implemented via additional series capacitors (notshown). The capacitor C0 forms with Ltap1, tap2, L0 and L01 . . . L0 n aparallel resonant circuit. If more inductors L01 . . . L0 n are used,any number of them can be coupled by switches Q01 . . . Q0 n to theresonant circuit. In any branches of Q0 n, L0 n additional seriesinductors and their switches can be implemented (not shown). Theseswitches bypass these series inductors, depending on their state. Suchextended inductor configurations can be required at high VCC′, highertransmitted power and/or multiple load units.

A load unit comprises a load resistor (Rload), a pickup inductor (Lsec)and often, but not necessarily a capacitor Csec, that tunes the loadunit's resonance frequency to match with the base station radiationfield. Alternatively, a parallel resonant circuit can also be used inthe load unit (not shown). The proper tuning to either series- orparallel resonance in the load unit is not critical, because theattenuation of Rload increases the bandwidth significantly.

All or any number of inductors Ltap1, Ltap2, L0, L01 . . . L0 n areimplemented as open coils to couple one or more load units to theresonant circuit of the base station. If several inductors Ltap1, Ltap2,L0, L01 . . . L0 n are used, they are positioned on a surface or onmultiple surfaces which are at an angle to each other. Is thecorresponding coil coupling switch Q01 . . . Q0 n closed, the current inthe corresponding inductance L01 . . . L0 n generates an inductionvector. In this manner, the induction field can be distributedarbitrarily on an area or a confined space. Advantageously, the radiatedinduction field is concentrated to locations where the load units areplaced. Thus, only those inductors L01 . . . L0 n are coupled to theresonant network, which transfer power to the load units. If only oneload unit is present, its coupling with the base station can be improvedby coupling more inductors L01 . . . L0 n to the resonant network.

The arrangement can be considered as a open transformator in the nearfield, thereby coupling factor (coupling k) varies with the geometricpositioning and with the distance from Ltap1, Ltap2, L0, L01 . . . L0 nto Lsec. The feedback includes a feedbackloop to couple the resonantnetwork to the input of the switches Qv1 and Qv2. The coil DR act as afilter in the supply line and reduces the harmonic distortions of thepush-pull stage (Qv1, Qv2). Substantially, DR acts as a current sourceto the tapping point, where a full wave rectified voltage waveform isdeveloped, whose mean is equal to VCC′. The minimum value of the voltageat the tapping point corresponds to the zero voltage crossing of thevoltage across C0. This zero voltage crossing representing the temporalminimum value of the tapping voltage of the coils Ltap1, Ltap2 triggersa frequency divider within the block Feedback. The inverted respectivelythe noninverted output of the frequency divider controls Qv1 or Qv2. Isfor example Qv1 closed and Qv2 open, a sinusoidal voltage half wave isgenerated at the entrance of the Feedback block, whose amplitude is halfthe amplitude of the voltage across C0. When this voltage reaches itsminimum, the frequency divider is triggered and the drive signals of Qv1and Qv2 change their states. Qv2 is closed and Qv1 is open. Again, asinusoidal voltage half-wave is generated, whose amplitude is half theamplitude of the voltage across C0. When this voltage reaches again itsminimum, the frequency divider is triggered and the drive signals of Qv1and Qv2 change again their states.

One or both drive signals from Qv1 respectively Qv2 or the triggersignal of the frequency divider within the block Feedback is compared asfist versus a reference frequency (fsoll) within the phase comparator(Phase Comp.). The error signal from the block Phase Comp. is filteredin the block Loopfilter. The filtered error signal controls the inductorL0 of the resonant network.

If the oscillator does not oscillate with fsoll, or changes theadmittance in the oscillator, an error signal is obtained at the outputof the phase (Phase Comp.). The filtered error signal controls theinductor L0 until the oscillator frequency (fist) becomes equal to thereference frequency (fsoll). In this manner, different inductor valuesin L01 . . . L0 n, changes in the load unit, component value changes andaltering coupling conditions are balanced and compensated. Theintegrating portion of the PLL controls the residual error always tozero. The PLL lock time is determined by the response time of L0 and thestability in the PLL. In the current invention L0 has a response time ofhalf a resonant circuit period. Thus, the PLL is dimensioned that itslock time becomes minimal possible after an unlock condition.

The blocks VCC control, Radicontrol and also their input and outputssignals behave identically to their eponymous in FIG. 3. This means thesame modes are implemented.

FIG. 7 shows a block diagram of a controlled large signal inductor(L0(control)) according to FIG. 6. Signal waveforms of FIG. 7 are shownin FIG. 8 and are used in the further description of its indices. Theoscillator is connected at the ports 1 and 2. As a simplification onlyone inductor Ltap (resulting inductor from Ltab1 and Ltab2) is shown inFIG. 7. Ltap form together with C0 a parallel resonant circuit. Theinductor LM is coupled to Ltap via coupling switches Q1 a and Q1 b. Thefirst limiting conditions of the switch control is given when thecoupling switches Q1 a and Q1 b are always open. Then, the resultingtotal inductivity is equal to Ltap and therefore maximal. Consequently,the resonance frequency is minimal. The second limiting conditions ofthe switch control is the one when the coupling switches Q1 a and Q1 bare always closed. Then the resulting total inductivity is equal to theparallel circuit of Ltap and LM and therefore minimal. Consequently, theresonance frequency is maximal. For part interval coupling of LM withLtap, any desired intermediate value between the two extreme values canbe set for the total inductivity by the control input Control. Theresonant circuit frequency is obtained as the average of the entireresonant circuit period.

The actual resonant circuit frequency changes within the resonantcircuit period between minimum and maximum respectively maximum andminimum based on the states of the two coupling switches. The resultinginductor current is sensed in block I-Sense and coupled to block SignDetect for sign determination. The inductor current sensing can comprisea current tapping. It is only important that the sign of the outputsignal has a defined phase relationship to the current I0 of theinductor L0. The resulting waveform is shown in A. The followingdifferentiator (Diff) derives A after the time and outputs its absolutevalue, see trace B. This results in positive pulses for each zerocrossing instants 0, T3 and T6 of the resonant circuit current I0. Thepulse width modulator (PWM) generates a sawtooth C (not shown in FIG. 6,since it is a signal inside the PWM modulator), which is synchronouswith the pulses B. This sawtooth trace C is compared versus the controlvalue Control inside the PWM block. Is Control lower than C, the PWMoutput D is high. Is Control greater than C, the PWM output D is low.Based on the known pulse width modulation (PWM) principle thePWM-modulator can be realized in various ways. It is only essential thatthe PWM signal D is triggered by B and the pulse width is controlled bythe input Control. The voltage across the Ltap is sensed in blockV-Sense and the following block Sign Detect determines its sign, seesignal J. The following differentiator (Diff) derives J after the timeand outputs its absolute value, see curve K. This results in positivepulses for each zero crossing of the resonant circuit voltage V0 (see V0zero crossing in FIG. 7). The timing of the signal pulses K coincidewith the maxima and minima of the current I0. The signal C is associatedwith A and D in Demux to generate the drive signals E′ and F′ for theswitches Q1 a and Q1 b. The switch Q1 a respectively Q1 b is closed(ON), if E′ respectively F′ is high. If E′ respectively F′ is low, thenQ1 a respectively Q1 b is open. Is A high, the signal D controls via E′the closing of switch Q1 a (see time instant T1). Is A low, the signal Dcontrols via F′ the closing of switch Q1 b see (time instant T4). Thetransition from high (ON) to low (OFF) in E′ and F′ is controlled bysignal K in conjunction with signal A, see time points V0 zero crossingin E′ and F′. Signal A selects the switch being controlled. Is A low,the switch Q1 a opens via E′ with K. Is A high, the switch Q1 b opensvia F′ with K. The ON to OFF transition control of the switches by usingK, halves the diode current flow interval comparing to control signalsthat are generated only by A and D (not shown in FIG. 7). The reduceddiode current flow interval results in smaller losses and prevents theformation of turn on transients in the diodes. There may still flow acurrent in the diodes. This process reduces in any case losses,especially if the resistance of the switches (Rdson) is sufficientlysmall. When E′ becomes high, the coupling switch Q1 a closes and itdevelops a half sinusoidal current wave in IM. During this interval, thetotal coupling switch voltage VQ1=VQ1 a+VQ1 b is zero, because bothcoupling switches are closed. If the current IM becomes zero at timeinstant T2, the stored energy in C0 generates a negative voltage step inthe internal diode of Q1 b and consequently the current flow interrupts.When F′ becomes high, the coupling switch Q1 b closes and it develops anegative half sinusoidal current wave in IM during the time intervalfrom T4 to T5. During this interval, the total coupling switch voltageVQ1 is zero, because both coupling switches are closed. If the currentIM becomes zero at time instant T5, the stored energy in C0 generates anegative voltage step in the internal diode of Q1 b and consequently thecurrent flow interrupts. The current flow interval of the internal diodeis limited to the interval from V0 zero crossing to T2 of the positivehalf wave of IM respectively T5 in the negative half wave of IM.

In a further embodiment of FIG. 6, L0 (control) is divided into two ormore parallel-controlled inductors. In that case each branch includes apartial inductor of L0 (control) and either one or two of switches (Q1 aand/or Q1 b). Is per partial inductor L0 (control) only one switchimplemented, then, in the second partial inductor is also only oneswitch implemented and they are driven that the combined current perinductors pair results in a waveform corresponding to the waveform IM inFIG. 8. Thus, this advantageously maintains the symmetry and keeps theharmonic components in the oscillator and in the radiated inductionfield minimal.

An oscillator also remains functional when a controlled resonant networkis used to control the frequency of a resonant circuit. According to afurther embodiment, FIG. 9 shows a parallel push-pull oscillator basedon the one in FIG. 6. The frequency control includes not only acontrolled inductance but at least one or more series circuits C01, L01. . . C0 n, L0 n, which are coupled via switches to the parallelresonant circuit (Ltap1, Ltap2 and C0). The switches are integrated herein block Multicoil Controller and they are used for frequency andinduction field control. The corresponding input signals are the outputof the Loopfilter (control) and coilselect from the block Radicontrol.The filtered error signal (control) controls one or more series circuitsC01, L01 . . . C0 n, L0 n of the resonant network.

If the oscillator does not oscillate with fsoll, or changes theadmittance in the oscillator, an error signal is obtained at the outputof the phase (Phase Comp.). The filtered error signal controls one ormore series circuits C01, L01 . . . C0 n, L0 n until the oscillatorfrequency (fist) becomes equal to the reference frequency (fsoll). Inthis manner different inductor values in L01 . . . L0 n, changes in theload unit, component value changes and altering coupling conditions arebalanced and compensated. The integrating portion of the PLL controlsthe residual error always to zero. The PLL lock time is determined bythe response time of the series circuit C01, L01 . . . C0 n, L0 n andthe stability in the PLL. In the current invention the series circuitC01, L01 . . . C0 n, L0 n has a response time of half a resonant circuitperiod. Thus, the PLL is dimensioned that its lock time becomes minimalpossible after an unlock condition.

The blocks VCC control, Radicontrol and also their input and outputssignals behave identically to their eponymous in FIG. 3. This means thesame modes are implemented.

All inductors (L01 . . . L0N) in the oscillator and alternatively alsoLtap1, Ltap2 can contribute to the power transmission from the basestation to the load units. The oscillator is functionally identical withFIG. 6. The description of FIG. 6 addresses implementation variantsconcerning additional inductor coupling to the parallel resonant circuit(Ltap1, Ltap2 and C0) and they apply without restrictions also to theFIG. 9. Functionally, the frequency control by means of series circuitsC01, L01 . . . C0 n, L0 n behaves substantially identical to theinductor control of FIG. 6.

The coupling switch control of the resonant network according to FIG. 9is shown in FIG. 10. The corresponding signals are shown in FIG. 11 andare used in the further description with their indices. The oscillatorcoupling ports are 1 and 2. As a simplification, the resulting totalinductance Ltap (from Ltap1 and Ltap2) is shown in FIG. 10. It formstogether with the capacitance C0 a parallel resonant circuit. One ormore series circuits C01, L01 . . . C0 n, L0 n are coupled via one ormore coupling switches Q01 . . . Q0 n with to Ltap. The first limitingcondition of the switch control is the one when all coupling switchesQ01 . . . Q01 n are always open. Then, the resulting total inductivityis equal to Ltap and therefore maximal. Consequently, the resonancefrequency is minimal. The second limiting condition of the switchcontrol is the one when the coupling switches Q01 . . . Q01 n are alwaysclosed. Then the resulting total inductivity is equal to the parallelcircuit of Ltap and the resulting inductivity of all series circuit C01,L01 . . . C0 n, L0 n and therefore minimal. Consequently, the resonancefrequency is maximal. The series circuits C01, L01 . . . C0 n, L0 n aredimensioned such that the series resonance frequency for permanentlyclosed switches Q01 . . . Q01 n is lower than the lowest networkfrequency respectively lowest oscillator frequency. In this case, theseries circuits act as inductors and thus can control the totalinductance of the entire network. For part interval coupling of C01, L01. . . C0 n, L0 n with Ltap, any desired intermediate value between thetwo extreme values can be set for the total inductivity by the controlinput control. The resonant circuit frequency is obtained as the averageof the entire resonant circuit period. The actual resonant circuitfrequency changes within the resonant circuit period according to thestates of the two coupling switches.

The resulting inductor current is sensed in block I-Sense and coupled toblock Sign Detect for sign determination. The inductor current sensingcan comprise a current tapping of I0. It is only important that the signof the output signal has a defined phase relationship to the current I0of the inductor Ltap. The resulting waveform is shown in A. Thefollowing differentiator (Diff) derives A after the time and outputs itsabsolute value, see trace B. This results in positive pulses for eachzero crossing instants 0, T3 and T6 of the resonant circuit current I0.The pulse width modulator (PWM) generates a sawtooth C (not shown inFIG. 10, since it is a signal inside the PWM modulator), which issynchronous with the pulses B. This sawtooth trace C is compared versusthe control value Control inside the PWM block. Is Control lower than C,the PWM output D is high. Is Control greater than C, the PWM output D islow. Based on the known pulse width modulation (PWM) principle thePWM-modulator can be realized in various ways. It is only essential thatthe PWM signal D is triggered by B and the pulse width is controlled bythe input Control. This pulse length corresponds to a controlled delaywhich starts at the trigger moment (signal B) and whose length iscontrolled by control.

The voltage across the Ltap is sensed in block V-Sense and the followingblock Sign Detect determines its sign, see signal G. The followingdifferentiator (Diff) derives G after the time and outputs its absolutevalue, see curve H. This results in positive pulses for each zerocrossing of the resonant circuit voltage V0 (see V0 zero crossing inFIG. 7). The timing of the signal pulses H coincide with the maxima andminima of the current I0. The signal H is associated with A and D inDemux to generate the drive signals E″ and F″ for the switches Q01 . . .Q0 n. The drive signals E″ and F″ are coupled via the AND gates A01 . .. A0 n to the corresponding coupling switches. E″ is identical to 01, 03. . . 0 n−1 and F″ is identical to 02, 04 . . . 0 n, if coilselectreleases all ANDs A01 . . . A0 n on E″ and F″. The correspondingselection signals coil 01 select . . . coil 0 n select are coupled bysignal A to the corresponding AND inputs. In this way, coil 01 select .. . coil 0 n select are allowed to change at arbitrary times. There isalways a well defined drive signal 01 . . . 0 n locked to the networkperiod. The switches Q01 . . . Q0 n are closed (ON), if E″ respectivelyF″ is high. If E″ respectively F″ is low, then, Q01 . . . Q0 n are open.Is A high, signal D controls via E″ the closing of switch Q01 (see timeinstant T1). Is A low, signal D controls via F″ the closing of switchQ02 (see time instant T4). The transition from high (ON) to low (OFF) inE″ and F″ is controlled by signal H in conjunction with signal A, seetime points V0 zero crossing in E″ and F″. Signal A selects the switchbeing controlled. Is A low, switch Q01 opens via E″ with H. Is A high,switch Q02 opens via F″ with H. The switches are always at least ahalf-period closed and thus bridge at least partially their internaldiodes. The reduced diode current flow interval results in smallerlosses and prevents the formation of turn on transients in the diodes.There may still flow a current in the diodes. This process reduces inany case losses, especially if the resistance of the switches (Rdson) issufficiently small. When E″ becomes high, the coupling switch Q01 closesand it develops a current wave in I01. This current wave represents acoupling interval which is twice as long as the interval from T1 to T3.During this whole coupling interval is the coupling switch voltage VQ01zero. At time instant T3 the total energy of the series circuit C01, L01is stored in C01. Further, the current I01 passes through its zero pointat time T3 and subsequently discharges the capacity C01 by a negativecurrent I01 in the interval from T3 to T5. Becomes the current I01 againzero at the time instant T5, the current flow interrupts until E″ thenext coupling interval of the series circuit C01, L01 and the rest ofthe resonant network releases. When F″ becomes high, the coupling switchQ02 closes and it develops a current wave in I02. This current waverepresents a coupling interval which is twice as long as the intervalfrom T4 to T6. During this whole coupling interval is the couplingswitch voltage VQ02 zero. At time instant T6 the total energy of theseries circuit C02, L02 is stored in C02. Further, the current I02passes through its zero point at time T6 and subsequently discharges thecapacity C02 by a negative current I02. This coupling half period isidentical to the interval from 0 to T2. Becomes the current I02 againzero at the time instant T2, the current flow interrupts until F″ thenext coupling interval of the series circuit C02, L02 and the rest ofthe resonant network releases. The resonance frequency of the overallnetwork changes depending on the coupling status of the series circuitsand the total network. In the intervals from 0 to T1 and T5 to T6, thetotal network period is determined by C0, Ltap, C02 and L02. At theintervals of T1 to T2 and T4 to T5, the overall network resonancefrequency is determined by C0, Ltap, C01, L01, C02 and L02. In theinterval from T2 to T4, the overall network period is determined by C0,Ltap, C01 and L01. This is valid, unless one takes into account the loadunit, otherwise a part of the resulting load influences in eachsub-interval the network additionally. By subsequent delayed turn on inthe coupling switch the coupling decreases the coupling period and thecircuit is driven less. Accordingly, the fewer changes the overallnetwork period respectively the overall frequency. Falls the couplingperiod below the half of the overall network period there is no moreoverlapping within the two coupling periods of C01, L01 and C02, L02.Advantageously, in this manner, with only one active element (with onlyone switch), an inductance (L01 . . . L0 n) is driven with a symmetricalcurrent wave. This results in less harmonics in the radiated inductionfield. However, lower harmonics are generated whenever a pair of seriescircuits (see C01, L01, Q01 and C02, L02, Q02) are coupled together tothe overall network in an anti-parallel configuration. The radiationfields in the two inductors L01 and L02 compensates in this case. Onecan easily recognize that even using one single switch (for example Q01)and one series circuit (for example C01, L01) a zero line symmetriccurrent waveform is generated. This leads for lower circuitimplementation costs and to a substantially purer radiated spectrum(smaller or less harmonic distortions) with respect to FIG. 6.

FIG. 12 shows a detailed circuit of a parallel push-pull large signalVCO according to the current invention. The oscillator correspondssubstantially to that of FIGS. 6 and 9. Ltap1, Ltap2 are in oneembodiment the ration coil (or at least part thereof) of the inductionvector. At each one terminal of the coupled inductors Ltap1, Ltap2 is aswitch Qv1 and Qv2 connected to ground. The block VCC control, hereimplemented as a switched mode power supply (SMPS), delivers theoperating voltage via DR to the center of Ltap1, Ltap2. The resonantcircuit is connected at terminals 1 and 2. This is in the simplest case,a controlled capacitor. In another embodiment, these are controlledcapacitors/inductors or controlled series circuits of the two.

RV9 and CV6 form a low pass, which dampens transients. QV7, DV3 and RV7form the voltage minimum detector whose reference voltage is filtered byCV5 and buffered by QV8. The reference voltage is determined by thevoltage divider RV6 and RV8. QV9 is only responsible for the startup inthe oscillator, after the startup operation QV9 remains alwaysconducting by RV5. The reference voltage tracks with the mean of theinput voltage by RV10 and CV7. This opens QV7 at the minimum of thevoltage across CV6 makes the switching operation in QV7 independent ofthe output voltage from the SMPS. The resulting pulses at the collectorQV7 trigger flip-flop FFV and let its outputs Q and IQ toggle.Consequently, the driver transistors QV3 . . . QV6 are always drivenwith opposite signals by CV1 and CV2 and let the switches Qv1 and Qv2also toggle inversely to each other. The components CV1, CV2, RV3 andRV4 ensure that both switch Qv1 and Qv2 remain open in case of a missingclock at the entrance of FFV. Because of this, RV3 and RV4 establish agrounded DC-level on both inputs of the drivers. In normal operation,DV1 and DV2 clamp the rectangular drive waveform to ground. This resultsin driver voltage levels from approximately 0 . . . 12V.

The startup of the circuit is done by the time constant of RV5, CV4. Thereference voltage across CV5 is equal to the voltage across CV7 at aninitialized startup and this remains QV7 in its open state. Becomes QV9conducting, the reference voltage across CV5 drops and QV7 turns on.This generates a first pulse, which toggles FVV and alters the drivesignals at Qv1 and Qv2. Once the half sine wave at the center of Ltap1,Ltap2 becomes again minimal, FFV triggers again. The switches Qv1, Qv2change their states and the oscillator starts. This digital control ofthe switches Qv1 and Qv2 guarantees the lowest losses, because theswitch state change occurs at the lowest voltage across the switches.This corresponds to zero voltage switching (ZVS).

The trigger signal from FFV is coupled as the signal fist to the phasecomparator (Phase Frequency Detector PFD). The comparison frequency(fsoll) is generated from a controlled frequency divider (RefDiv) whoseinput clock is a constant reference frequency (fref), for example 32MHz. The PFD is equivalent to the known circuit of the CMOS deviceCD4046. In contrast to CD4046, here, the PFD-output signals are coupledindependently via the resistors RP1 . . . RP4 and a current mirror QV8,QV10 respectively QV7, QV9 to the loop filter CP1, CP2 and RP7.Transconductance amplifier QV11 generates proportional to the controlvoltage developed across the loop filter a control current for thecontrolled admittances. The PLL-loop gain is altered through the analogswitch SW by means of RP2 and RP3. This changes the charge pump currentin QV9 and QV10, which flows into the loop filter.

FIG. 13 shows a detailed circuit of a controlled capacitor according toFIG. 4. The dashed lines include parts of the block diagram. Theoscillator coupling terminals are 1 and 2. In the further descriptionwaveforms of FIG. 5 are referenced with their indices. CM is designed asa series connection of two capacitors CM1 and CM2, which is coupled toQ1 a and Q1 b to C0. The switch control circuit operates at a supplyvoltage +12′. An opto coupler transmits the control information(Control) isolated, because the switch control circuit operating on+12′, shall be electrically isolated from the rest of the network. Thecapacitors C1 and C2 sense the positive voltage half-wave of V0 andcouple it via R1 to the base of transistor Q10. Q10 acts as aquasi-zero-voltage detector of the positive half wave. D1 and D3accelerate the turn off in Q10 as soon as the voltage across C2 fallsbelow the threshold voltage of Q10. The capacitive voltage divider C1,C2 prevents DC coupling with V0. In addition, the components R1, C1 andC2 may provide a phase advance. To this end, these components are sodimensioned that the phase advance value is at least as large as theentire signal delay of the control circuit. This allows the use of theentire control range of the circuit. The components of Q9, D2, D4, R3,R4, C3 and C4 operate identically to the components of Q10, beingresponsible for the negative half wave of V0. The two D flip-flop FF1,FF2 and the NAND gate N1 detect the two positive edges of the outputsignals of the zero-voltage detectors Q9 and Q10. This sets alternatelythe output Q of a flip-flop (FF1 respectively FF2) to high, once it istriggered on the clock CK. This state is immediately reverted by thecorresponding reset input after the propagation time of N1. This resultsin voltage pulses at the output of N1 at each zero crossing of V0(waveform B) consequently, this closes Q2 completely during therepresentative short time interval. This symmetrical implementationguarantees identical behavior with respect to the two voltage half-wavesof V0. In addition, Q2 is coupled with the voltage V0 only during thebrief moment of the trigger event. Q7, Q8, R6 and R5 mirror a controlcurrent, defined by the control variable control, which flows into thecapacitor C5. The result is a sawtooth voltage on C5 (curve C), whosetiming of its lowest level fall together with the control pulses of Q2.This sawtooth triggers FF5 via the input CK as soon as its switchingthreshold is reached. This time can be controlled via the control inputcontrol, because the slope of the sawtooth voltage changes by thecollector current in Q8 and thus varies with control. FF5, FF6, N3 andN4 form the first part of the Demux. FF5 is configured with pulses onthe Set (S) respectively Reset (R) inputs that the sawtooth alwaysbecomes responsible for the corresponding switch, which is responsiblefor the corresponding half-wave. That represents a switching signalselection by signal A. Appears V0 with the its positive half wave, D2 isconducting. The collector of Q9 is high and triggers FF2 on CK. Theshort appearing high pulse on output Q of FF2 set the output Q of FF5via its set (S) input. The low at output IQ of FF5 forces a high on theoutput of N4, which closes Q1 a by the drivers Q3, Q4. This is theinitial condition for the controlled decoupling of CM1 and CM2 duringthe positive half wave of V0. If the switching threshold of the PWMmodulator is reached (the voltage across C5 is equal to the threshold ofFF5), FF5 toggles: Q is low and IQ is high. IQ of FF6 is also high, thusthe output of N4 switches to low, consequently the coupling switch Q1 ais opened by the drivers Q3, Q4. A similar situation happens during thenegative half wave of V0. FF5 is reset via the R input by FF1 when D1 isconducting and FF1 triggers on the clock CK. This forces a high at theoutput of N3, which closes Q1 b by the drivers Q5, Q6. This is theinitial condition for the controlled decoupling of CM1 and CM2 duringthe negative half wave of V0. If the switching threshold of the PWMmodulator is reached (the voltage across C5 is equal to the threshold ofFF5), FF5 toggles: Q is high and IQ is low. Q of FF6 is also high, thusthe output of N3 switches to low, consequently the coupling switch Q1 bis opened by the drivers Q5, Q6. The closing of the switches isestablished by output of N1, which triggers FF6. Consequently theoutputs of FF6 become equal to those of FF5. This forces N3 respectivelyN4, which was previously at a low output low level to a high outputlevel. This ensures that from this point in time both switches Q1 a andQ1 b remain closed until the PWM modulator opens again one of the twoswitches.

After the circuit start-up, the flip-flop FF6 resynchronizes itselfafter a maximum time interval during one resonant circuit period. Then,always at least one of the coupling switches Q1 a respectively Q1 b hasbeen opened and a clock pulse is generated for FF6.

FIG. 14 shows a detailed circuit of a controlled series of a capacitorand inductor according to the block diagram of FIG. 10. The dashed linesinclude parts of the block diagram. The oscillator coupling terminalsare 1 and 2. In the further description waveforms of FIG. 11 arereferenced with their indices. L01, C01, L02, C02, L03, C03 and L04, C04are coupled to C0 via Q1 . . . Q4. The necessary coupling switch controlsignals 01 . . . 04, are coupled galvanically isolated from the outputsof the NAND gates N3 . . . N6 via the transformers T1 . . . T4 to thecoupling switches Q01 . . . Q04. Alternatively, optocouplers are usedwith subsequent drivers (not shown). The NANDs N3 . . . N6 form themultiplexer, which selects the switch control outputs of flip-flops FF5and FF6 dependent on the through FF7 and FF8 latched selection signalscoil 01 select and coil 02 select. Is coil 01 select respectively coil02 select high, the corresponding NANDs (N3, N4 respectively N5, N6) aretransparent and link the outputs of FF5 and FF6 to the correspondingcoupling switch pair (Q01, Q02 respectively Q03, Q04). In a furtherembodiment, all four inputs of the NANDs N3 . . . N6 can beindependently selected by 4 control lines coil 01 select . . . coil 04select (not shown in FIG. 12). The switch control circuit operates at asupply voltage +12 with arbitrary reference potential. The inductor L1sense the positive current half-wave of I0 and couples it by means ofthe shunt resistor R13 via R1 and C2 to the base of transistor Q10. Q10acts as a quasi zero-current detector of the positive half wave. D1 andD3 accelerate the turn off in Q10 as soon as the current in C2 changesits sign. The capacitor C2 prevents DC coupling with I0. In anotherembodiment, the current sense network is designed such that a phaseadvance results. In this manner the entire control range can beexploited. The components of Q9, D2, D4, R3, C4 and L2 operateidentically to the components of Q10, here, they are responsible for thenegative current half wave of I0. The two D flip-flop FF1, FF2 and theNAND gate N1 detected the two positive edges of the output signals ofthe zero-current detectors Q9 and Q10. This sets alternately the outputQ of a flip-flop (FF1 respectively FF2) to high, once it is triggered onthe clock CK. This state is immediately reverted by the correspondingreset input after the propagation time of N1. This results in voltagepulses at the output of N1 at each zero crossing of I0 (waveform B),this closes Q2 completely for the representative short time interval.This symmetrical implementation guarantees identical behavior withrespect to the two current half-waves of I0. In addition, Q2 is coupledwith the current I0 only during the brief moment of the trigger event.The loopfilter control current, inserted over terminal 3 (control),charges capacitor C5 by the current mirror Q7, Q8, R6 and R5. The resultis a sawtooth voltage on C5 (curve C), whose timing of its lowest levelfall together with the control pulses of Q2. This sawtooth triggers FF5via the input CK as soon as its switching threshold is reached. Thistime instant can be controlled via the control input control, becausethe slope of the sawtooth voltage changes by the collector current in Q8and thus varies with control. FF5, FF6, N3 and N4 form the Demux. FF5 isconfigured with pulses on the Set (S) respectively Reset (R) inputs thatthe sawtooth always becomes responsible for the corresponding switch,which is responsible for the corresponding half-wave. That represents aswitching signal selection by signal A. Appears 10 with its positivehalf wave, D2 is conducting. The collector of Q9 is high and triggersFF2 on CK. The short appearing high pulse on output Q of FF2 sets theoutput Q of FF5 via its set (S) input. The high at output Q of FF5forces a low on the outputs of N4 and N6, which opens Q1 and Q3 by thedrivers Q3, Q4 respectively Q15, Q16. This is the initial condition forthe controlled coupling of the series circuits L01, C01 and L03, C03during the positive half wave of I0. If the switching threshold of thePWM modulator is reached (the voltage across C5 is equal to thethreshold of FF5), FF5 toggles and its output Q is low. IQ of FF6 isalso high, thus the outputs of N4 and N6 switch to high, consequentlythe coupling switches Q01 and Q03 are closed by the drivers Q3, Q4respectively Q15, Q16 if coil 01 select and coil 02 select are high aswell. A similar situation happens during the negative half wave of I0.The inverted output IQ of FF5 is set via the reset (R) input by FF1 whenD1 is conducting and FF1 triggers on the clock CK. This forces a low onthe outputs of N3 and N5, which opens Q02 and Q04 by the drivers Q5, Q6respectively Q13, Q14. This is the initial condition for the controlledcoupling of the series circuits L02, C02 and L04, C04 during thenegative half wave of I0. If the switching threshold of the PWMmodulator is reached (the voltage across C5 is equal to the threshold ofFF5), FF5 toggles and its inverted output IQ is low. Q of FF6 is alsohigh, thus the outputs of N3 and N5 switch to high, consequently thecoupling switch Q02 and Q04 are closed by the drivers Q5, Q6respectively Q13, Q14 if coil 01 select and coil 02 select are high aswell.

The opening of the switches is controlled by Vsense and Diff. For this,the zero crossing of the voltage V0 is detected by a zero voltage sensoraccording to FIG. 13. The blocks V-Sense and Diff with all itscomponents are identical in FIGS. 13 and 14 and are therefore notfurther explained. The pulses H at the output of N2 signal the moment atwhich the corresponding switches shall open again. This “turn-offcommand” triggers FF6 on the Clock input. Consequently the outputs ofFF6 become equal to those of FF5. For the positive current half wave ofI0 remain N4 and N6 on a high output level, N3 and N5 become low ontheir outputs. For the negative half wave of I0 remain N3 and N5 on ahigh output level and N4 and N6 become low on their outputs. Thus fromthis point (V0 zero crossing) only one of the switches Q01, Q02respectively Q03, Q04 remain continuously closed for the correspondinghalf-wave of I0. Therefore, the decoupling of L01, C01 . . . L04, C04 isonly determined by the integrated diodes in Q01 . . . Q04. A phaseadvance in the block V-Sense might no longer be absolutely necessary.The important boundary condition that the switches Q01 . . . Q04 mustopen before the internal diode opens must be strictly adhered. Theinverted states of coil 01 select and coil 02 select are latched in theflip-flops N7 and N8 by the PWM synchronisation pulse (output of N1). Inthis manner, the number of operational series resonant circuits L01, C01. . . L04, C04 in the resonant network is selected arbitrarily with atemporal resolution of half a network period.

The advantage of this differentiation (edge detection using flip-flops)is the resilience to transients, which affect only the functiondisruptively when the total voltage (V0) changes its sign. This makesthe “turn-off command” of the switches Q01 . . . Q04 insensitive tonetwork transients. After the circuit start-up, the flip-flop FF6resynchronizes itself after a maximum time interval of one resonantcircuit period.

In a further embodiment of the invention, the capacitors C01 . . . C04are not implemented and replaced with bridges in the circuit of FIG. 14(not shown). Then, the VCO of the embodiment corresponds to the multiinductor implementation in FIGS. 7 and 8.

The Controller in FIG. 12 generates pcont coilselect (containing thesignals coil 01 select and coil 02 select) and fsoll according to inputdata, which are contained in radiationdata. Loopselect is generated fromthe data coilselect.

In another embodiment, loopselect is generated by using coilselect andRefDivvalue. RefDiwalue divides fref for generating fsoll.

In one embodiment, RefDiwalue remains constant in time and generates adiscrete VCO spectral line (measured according to EN300330). TheController controls this spectral line by Refdivvalue and fsollaccording to data which are included in radicontrol. Fsoll is thereinconstant during an energy transfer interval. An energy transfer intervalis a time interval which is characterized by a radiated induction vector(measured current in a radiating coil) developing more than 1% . . . 10%of its maximal amplitude.

In another embodiment, RefDivvalue changes in time and distributes aninduction vector (measured current in a radiation coil) to at least twospectral positions.

In another embodiment, RefDivvalue updates fsoll quickly and distributesan induction vector (measured current in a radiating coil) over severalspectral lines or a spectral range (measured according to EN300330).

A characteristic feature of all embodiments is that the update rate infsoll is all the time slower than the lock time of the PLL-control loop.In one embodiment, the Controller generates within a minimum and maximumbounded range a RefDiwalue which continuously changes randomly in time.These values are generated by a pseudo random sequence (PRS), and formin at least one part spectrum of the induction vector spectrum (measuredcurrent in a radiating coil) a sine(x)/x.

In one embodiment, the Controller generates within a minimum and maximumbounded range a RefDivvalue which continuously changes in a sweepingmanner in time. A sweep sequence is characterized in that the outputvalues periodically change from a minimum value to the maximum value andthereupon from a maximum to the minimum value and so on. FIG. 15 a showsa period of such a sweep sequence (RefDivvalue(t)), wherein theresidence time (time interval, where RefDivvalue remains constant) onthe time axis (t) changes dynamically depending on RefDivvalue. Forexample, a residence time of one time unit is used for the min and maxvalues and another residence time of five time units is used for theaverage of min and max. Furthermore, in one embodiment, the step valuesize of RefDivvalue is not constant. By means of this two describedmethods, as shown in FIG. 15 a, a cubic function in RefDiwalue(t) can beapproximated. Such a function produces a uniform spectrum, and ischaracterized by approximately the same maximum value IPI_(max)(f) (seeFIG. 15 b).

In a further embodiment not all possible RefDivvalues are assigned toform the function RefDiwalue(t). In this manner, as shown in FIG. 16 a,the value of “blacklisted RefDivvalue” is excluded in the sweepsequence. Consequently, there is no frequency fsoll generated at thepoint fref divided by “blacklisted RefDivvalue”. Therefore, no spectralcomponent at the corresponding VCO frequency position is generated. Inthis way, any desired notches in IPI_(max)(f) can be produced (see FIG.16 b). It is clear that this method applies to an arbitraryRefDivvalue(t) sequence, whether it is a sweep sequence or a PRS.

In one embodiment, the Controller of FIG. 12 generates pcont—and fsollvalues, which give an arbitrary spectral distribution including notches(for example according to EN300330, REC7003 and ITU-RSM2123). Pcont iseither determined by the power control between the base station and loadunit, or if necessary by limiting maximum values as mandated by thesestandards. Advantageously, a memory stores appropriate RefDivvalues(min, max, and one or more blacklist RefDivvalue) as parameters.Further, a memory stores parameters for Pcont and parameters whichdetermine the residence time of RefDivvalue(t).

The Controller can be implemented as a discrete digital network, anintegrated logic circuit (PLD, FPGA) or be embodied as amicrocontroller. Further, it has long been state of the art, thatRAM/ROM memories (Memory) are also integrated into integrated circuits(PLD, FPGA and microcontroller with built-in memory). It is also stateof the art that data input-/output interfaces (for example pcont,radiationdata, coilselect and Refdivvalue) are implemented by serial- orparallel busses. In this manner, the Controller implementation canrealized very compact, inexpensive and/or also according various otherrequirements.

All LC-based AC-generators and all existing LC-oscillator variants canbe constructed with a controlled capacitor and/or a controlled inductorand/or a controlled combination circuits of the two. The term “L” standsfor inductance respectively coil, and the term “C” stands forcapacitance respectively capacitor. The simplest embodiment of thepresent invention represents an AC voltage generator with a tuneableresonant network. In this embodiment, one simply eliminates the Feedbackblock in the FIGS. 3, 6 and 9.

The switches Qv1 and Qv2 are driven by an arbitrary external signal. Theindependent (decoupled from the oscillator) control signals (V1 and V2)provide a higher freedom in the generator design. This can beinteresting if exact resonance tuning can be neglected or the bandwidthof the resonant circuit is very high.

For the functioning of an oscillators, it does not matter, whether thefeedback (Feedback) in the oscillator is implemented by a transformer oran inductive-respectively a capacitive 3-point circuit.

It is clear that the method according to the present invention appliesdirectly to full-bridge circuits, regardless of whether they areimplemented as output stages or oscillators.

Another embodiment of the present invention combines multiple VCOsimplemented according to the current invention, which operate in acombined manner to induction coils.

Since the current phasing, respectively, the voltage phasing in theresonant circuit of the VCOs are well defined, induction fieldsuperposition is particularly easy to realise using the currentinvention.

The invention claimed is:
 1. An alternation voltage- or an alternationcurrent generator whose load network can be tuned in frequencycomprising; a) at least one first switch controlled by an alternationvoltage or an alternation current, b) at least a first inductor and atleast a first capacitor which form a load resonant circuit of said firstswitch, c) at least one reactive component coupled by at least onesecond switch to said load resonant circuit, wherein said reactivecomponent is charged and discharged by said load resonant circuit andsaid second switch alters its state at least once within a load resonantcircuit period, d) a power control circuit responsive to a amplitudecontrol signal to control at least one of the voltage- or currentamplitude in said load resonant circuit, and e) a control circuit tocontrol said second switch responsive to a non-zero threshold level of asign change of at least one of a voltage or a current in said loadresonant circuit, wherein at least one of the load resonant circuitimpedance or load resonant circuit admittance is controlledsubstantially symmetrically with respect to at least one of the positiveand negative voltage- or current half wave of said load resonantcircuit.
 2. An alternation voltage- or an alternation current generatoraccording to claim 1, wherein said control circuit generates a digitalPWM drive signal for said second switch and said control circuit isfurther responsive to a frequency control signal to control the dutycycle of said digital PWM drive signal.
 3. An alternation voltage or analternation current generator according to claim 1, wherein said firstswitch comprising at least two switches which form one of a half- or afull bridge or a push pull circuit and said second switch comprising atleast two switches.
 4. An alternation voltage or an alternation currentgenerator according to claim 1, wherein said reactive component is atleast one of; c1) a second inductor, c2) a second capacitor, and c3) aserial circuit formed by a third inductor and a third capacitor.
 5. AnOscillator whose frequency is controlled by input a frequency controlsignal comprising; a) at least one first switch, b) at least a firstinductor and at least a first capacitor which form a resonant circuit,wherein said first switch is coupled with its input and output in apositive feedback manner to said resonant circuit to form an oscillatorto generate an alternation voltage or current in said resonant circuit,c) at least one reactive component coupled by at least one second switchto said resonant circuit, wherein said reactive component is charged anddischarged by said resonant circuit and said second switch alters itsstate at least once within a resonant circuit period, d) a power controlcircuit responsive to an amplitude control signal to control at leastone of the voltage- or current amplitude in said resonant circuit, e) acontrol circuit that controls said second switch responsive to saidfrequency control signal, wherein at least one of said resonant circuitimpedance or said resonant circuit admittance is controlledsubstantially symmetrically with respect to at least one of the positiveand negative voltage- or current half wave of said resonant circuit. 6.An oscillator according to claim 5, wherein said oscillator comprising adrive circuit which is responsive to a non-zero threshold level of asign change of at least one of a voltage or a current in said resonantcircuit to generate a digital drive signal for said first switch.
 7. Anoscillator according to claim 5, wherein said first switch comprising atleast two switches which form one of a half- or a full bridge or a pushpull circuit and said second switch comprising at least two switches. 8.An oscillator according to claim 5, wherein said frequency controlsignal is generated in a phase locked loop (PLL) by a comparison of oneof a resonant circuit voltage or current versus a reference signal. 9.An oscillator according to claim 5, wherein said control circuit tocontrol said second switch is responsive to a non-zero threshold levelof a sign change in at least one of a voltage or a current in saidresonant circuit.
 10. An oscillator according to claim 9, wherein saidcontrol circuit detects at least one of a voltage- or current half waveof said resonant circuit, wherein said threshold level is substantiallyidentical for both voltage- and current half waves of the resonantcircuit.
 11. An oscillator according to claim 5, wherein said reactivecomponent is at least one of; c1) a second inductor, c2) a secondcapacitor, and c3) a serial circuit formed by a third inductor and athird capacitor.
 12. An oscillator according to claim 6, wherein saiddrive circuit detects at least one of a voltage- or current half wave ofsaid resonant circuit, wherein said threshold level is substantiallyidentical for both voltage- and current half waves of the resonantcircuit.